Methods and apparatus for determining an interferometric angle to a target in body coordinates

ABSTRACT

A method for processing radar return data to determine a physical angle, in aircraft body coordinates to a target, is disclosed. The radar return data includes a phase difference between radar return data received at an ambiguous radar channel and a left radar channel, a phase difference between radar return data received at a right radar channel and an ambiguous radar channel, and a phase difference between radar return data received at a right radar channel and a left radar channel. The method includes adjusting a phase bias for the three phase differences, resolving phase ambiguities between the three phase differences to provide a signal, and filtering the signal to provide a physical angle to the target in aircraft body coordinates.

BACKGROUND OF THE INVENTION

[0001] This invention relates generally to radar systems, and morespecifically to a radar system which is capable of synchronization witha digital elevation map (DEM) to accurately determine a location.

[0002] The proper navigation of an aircraft in all phases of its flightis based to a large extent upon the ability to determine the terrain andposition over which the aircraft is passing. In this regard,instrumentation, such as radar systems, and altimeters in combinationwith the use of accurate electronic terrain maps, which provide theheight of objects on a map, aid in the flight path of the aircraft.Electronic terrain maps are well known and are presently used to assistin the navigation of aircraft.

[0003] Pulse radar altimeters demonstrate superior altitude accuracy dueto their inherent leading edge return signal tracking capability. Thepulse radar altimeter transmits a pulse of radio frequency (RF) energy,and a return echo is received and tracked using a tracking system. Theinterval of time between signal bursts of a radar system is called thepulse repetition interval (PRI). The frequency of bursts is called thepulse repetition frequency (PRF) and is the reciprocal of PRI.

[0004]FIG. 1 shows an aircraft 2 with the Doppler effect illustrated byisodops as a result of selection by the use of Doppler filters. The areabetween the isodops of the Doppler configuration will be referred to asswaths. The Doppler filter, and resulting isodops are well known in thisarea of technology and will not be explained in any further detail.Further, the aircraft 2 in the specification will be assumed to have avertical velocity of zero. As is known, if a vertical velocity exists,the median 8 of the Doppler effect will shift depending on the verticalvelocity. If the aircraft 2 has a vertical velocity in a downwarddirection, the median of the Doppler would shift to the right of thefigure. If the aircraft 2 has a vertical velocity in an upwarddirection, the Doppler would shift to the left of the figure. Again, itwill be assumed in the entirety of the specification that the verticalvelocity is zero for the ease of description. However, it is known thata vertical velocity almost always exists.

[0005] Radar illuminates a ground patch bounded by the antenna beam 10from an aircraft 2. FIG. 1a shows a top view of the beam 10 along withthe Doppler effect and FIG. 1b shows the transmission of the beam 10from a side view. To scan a particular area, range gates are used tofurther partition the swath created by the Doppler filter. To scan acertain Doppler swath, many radar range gates operate in parallel. Withthe range to each partitioned area determined, a record is generatedrepresenting the contour of the terrain below the flight path. Theelectronic maps are used with the contour recording to determine theaircraft's position on the electronic map. This system is extremelycomplex with all the components involved as well as the number ofmultiple range gates that are required to cover a terrain area. As aresult, the computations required for this system are very extensive.

[0006] In addition to the complexity, the precision and accuracy of thedistance to a particular ground area or object has never been attainedusing an airborne radar processor.

BRIEF SUMMARY OF THE INVENTION

[0007] In one aspect, a method for processing radar return data todetermine a physical angle, in aircraft body coordinates to a target isprovided. The radar return data includes a phase difference betweenradar return data received at an ambiguous radar channel and a leftradar channel, a phase difference between radar return data received ata right radar channel and an ambiguous radar channel, and a phasedifference between radar return data received at a right radar channeland a left radar channel. The method comprises adjusting a phase biasfor the three phase differences, resolving phase ambiguities between thethree phase differences to provide a signal, and filtering the signal toprovide a physical angle to the target in aircraft body coordinates.

[0008] In another aspect, a processor is provided which is configured toresolve phase ambiguities between multiple received phase differencesignals and determine a physical angle in aircraft body coordinates to atarget based upon the resolved phase ambiguities, the phase differencesignals having been determined based upon radar return data received ateach of an ambiguous radar channel, a left radar channel, and a rightradar channel.

[0009] In still another aspect, a radar signal processing circuit isprovided. The circuit comprises a radar gate correlation circuitconfigured to sample radar return data from left, right, and ambiguousradar channels at a sampling rate, a correlation bass pass filterconfigured to stretch the sampled radar return data to a continuous wave(CW) signal, and a mixer configured to down sample an in-phase componentand a quadrature component of the CW signal to a doppler frequency Theradar signal processing circuit further comprises a band pass filtercentered on the doppler frequency, a phase processor configured toreceive processed radar return data from the band pass filter, and aprocessing unit configured to receive the three phase differences. Thephase processor is configured to determine a phase difference betweenradar return data from an ambiguous channel and a left channel, a phasedifference between radar return data from an right channel and theambiguous channel, and a phase difference between radar return data fromthe right channel and the left channel. The processing unit isconfigured to receive the three phase differences, adjust a phase biasfor the three phase differences, resolve phase ambiguities between thethree phase differences to provide a signal, and filtering the signal toprovide a physical angle to a target in aircraft body coordinates.

BRIEF DESCRIPTION OF THE DRAWINGS

[0010]FIG. 1a is a diagram illustrating swaths made by a radar.

[0011]FIG. 1b is a diagram illustrating a radar transmit pattern.

[0012]FIG. 2 is an illustration of radar signal waveforms over time.

[0013]FIG. 3 is a diagram illustrating radar signals being received bythree antennas.

[0014]FIG. 4 is a diagram illustrating a body coordinate system.

[0015]FIG. 5 is a diagram illustrating a doppler coordinate system withrespect to the body coordinate system of FIG. 4

[0016]FIG. 6 is a block diagram of a radar signal processing system.

[0017]FIG. 7 is a block diagram of a digital sampling and filteringsection.

[0018]FIG. 8 is a block diagram of a correlation band pass filter.

[0019]FIG. 9 is a block diagram of a in-phase/quadrature mixer.

[0020]FIG. 10 is a block diagram of an all pass filter network forin-phase and quadrature components of a signal, within the mixer of FIG.8.

[0021]FIG. 11 is a diagram of a second order all pass filter.

[0022]FIG. 12 is a block diagram of a swath band pass filter.

[0023]FIG. 13 is a block diagram of a filter coefficients processor.

[0024]FIG. 14 is a velocity vector diagram.

[0025]FIG. 15 is a block diagram of a phase processor including threephase detectors.

[0026]FIG. 16 is a block diagram of one phase detector from FIG. 15.

[0027]FIG. 17 is a block diagram of an interferometric angle resolver.

[0028]FIG. 18 is a chart illustrating varying electrical phasedifferences between three antenna pairings.

[0029]FIG. 19 is a block diagram which illustrates inputs to a bodycoordinate processor.

[0030]FIG. 20 is a block diagram of the body coordinate processor ofFIG. 19.

[0031]FIG. 21 is an illustration of the derivation of a doppler circle.

[0032]FIG. 22 is an illustration of the derivation of an interferometriccircle.

[0033]FIG. 23 is a diagram illustrating barker coded transmit andreceive pulses.

[0034]FIG. 24 is a block diagram illustrating inputs to and outputs froma range verification processor

[0035]FIG. 25 is a flowchart illustrating a range verification method.

DETAILED DESCRIPTION OF THE INVENTION

[0036] There is herein described a combination Dopplerradar/interferometer to navigate an aircraft 2 with respect to terrainfeatures below aircraft 2. As used herein, aircraft is used to identifyall flight platforms which may incorporate a radar system, including,but not limited to, jets, airplanes, unmanned aerial vehicles, missiles,and guided weapons. The radar also functions with an electronic map,sometimes referred to herein as a digital elevation map (DEM), indetermining a position of aircraft 2. In addition to determining analtitude of aircraft 2, an XYZ location of the nearest object toaircraft 2 on the ground, with respect to aircraft 2 in a certainterrain area can be determined. As aircraft 2 is flying over terrain asshown in FIGS. 1a and 1 b, it is important to determine a position ofaircraft 2 in accordance with a map. A Doppler filter and range gate areused with a transmitted beam 10 from a transmit antenna.

[0037] In a general altitude range tracking radar, range is measured andindicated by measuring the time for transmitted energy to be reflectedfrom the surface and returned. With reference to FIG. 2, a radartransmitter repeatedly sends out bursts of electromagnetic energy at apredetermined repetition rate from an antenna, as indicated by transmitpulse 20. Following a time delay which is a function of the aircraftaltitude, a ground return pulse 22 is received by a receiving antennafeeding a receiver. A range gate 30 is utilized by the tracking radar toview at least a portion of ground return 22.

[0038] Referring to FIG. 3, three receive antennas, antenna R (right)42, Antenna L (left) 44, and an ambiguous antenna (Ant Amb) 46, are usedto receive information. Along with the three antennas, three processingchannels, referred to below as left, right and ambiguous respectively,each include a receiver, a data acquisition device, range gate, and afilter. Use of the three antenna system, along with the processingdescribed herein, provides a solution to ambiguous detected angle of thenearest object. The ambiguous detected angle is due to the spacing ofthe antennas being greater than the transmitted RF frequency wavelength.By receiving three returns, the processing system is able to determinean umambiguous location of the nearest object on the ground, which inturn is utilized to locate position of aircraft 2 in body coordinates.Body coordinates are typically preferable than positioning as determinedby known systems, as those systems determine position as if the bodyaircraft 2 is aligned with the line of flight. As aircraft 2 is prone topitch, roll, and yaw, the body of aircraft 2 is not necessarily alignedwith the line of flight.

[0039] In an exemplary illustration, antenna R 42, along with processingsystems (described below) will provide a course range search whichroughly determines the range to the nearest point 48 in swath 12 (shownin FIG. 1) before aircraft 2 has passed over from swath 14 into swath12. Determination of the nearest point 48 is performed by a widebandwidth, high speed track loop which quickly determines the range tonearest point 48 in swath area 12. Nearest point 48 provides a startingpoint for a tracking loop using antenna L 44 and ambiguous antenna 46.The track loop controls the range gate to track returns from a transmitantenna. A narrow bandwidth, high precision processor is used to setrange gates for antenna L 44 and ambiguous antenna 46 to an exact rangeof nearest point 48 based on the previous course range determination.The operation of the three receive antennas and associated processingchannels provides a quick and accurate setting of a range gate on thenearest object in the Doppler swath 14 directly below aircraft 2 so thata phase difference can be measured and along with the known separations50 amongst the three antennas, a crosstrack distance to the object 48 isdetermined. The crosstrack distance is the distance, horizontal andperpendicular to the body coordinates of aircraft 2, to object 48.

[0040]FIG. 3 shows a view with aircraft 2 going into the Figure. Duringthe phase comparison portion of the time interval, the Doppler filtersof the left, right and ambiguous channels are set to select a swath 14(shown in FIG. 1) below aircraft 2. Further, both range gates are set ata range directly on the nearest object 48 as previously determined. Fromthis range, antenna R 42 receives a signal from object 48 at a distanceof R1, ambiguous antenna 46 receives a signal from the object 48 at adistance of RA, and antenna L 44 receives the signal from object 48 at adistance of R where the distance difference is a function of the antennaseparation 50 between and amongst the three antennas. A phase processor(described below) compares the phase difference between R1 and RA, R2and RA, and R1 and R2 once the return signals are received. Asillustrated in the Figure, the exact range differences (R2−R1), (RA−R1),and (R2−RA) are from phase differences and simple trigonometry relationsare used to determine the exact crosstrack distance to the object 48 inaircraft body coordinates.

[0041] As illustrated in FIG. 3, after the range differences (R2−R1),(RA−R1), and (R2−RA) are determined and knowing the antenna separations50, and measured range R1, then the crosstrack distance (Y) and verticaldistance (Z) can also be computed in aircraft body coordinates. It isimportant that the precise location of nearest object 48 in each swathis determined so correlation can be made with the electronic maps whichwill accurately locate the aircraft 2 on the electronic map. Forexample, at typical high speed aircraft cruising velocities, a radar,configured with reasonably sized Doppler filters, has swath widths ofapproximately 10 feet at 5000 feet altitude. The resulting incidenceangle formed by the intersection of R1 and a vertical line 27 will thenbe on the order of less than 3 degrees. Basic trigonometry relationsshow that even with a typical error (for example 1%) on the radar rangegate measured distance R1, (50 feet at 5000 feet altitude), knowing theprecise antenna separation 50, and precise range differences (R2−R1),(RA−R1), and (R2−RA), the crosstrack distance (Y) will be precise due tothe very small incidence angle encountered.

[0042]FIG. 4 illustrates a body coordinate system. The body coordinatesystem, is the coordinate system with respect to aircraft body 2. Anx-axis, Xm is an axis which passes through a nose of aircraft body 2. Ay-axis, Ym, is an axis which is 90 degrees from Xm and is positive tothe right of aircraft body 2. A z-axis, Zm, is an axis which is 90degrees from both Xm and Ym and perpendicular to a bottom of aircraftbody 2. With respect to aircraft maneuvering, a positive roll is a dropof the right wing, a positive pitch is a nose up, and a positive yaw isthe nose to the right, all with respect to a line of flight.

[0043] It is known that aircraft do not typically fly in alignment withthe aircraft body coordinates. Such a flight path is sometimes referredto as a line of flight. Therefore an aircraft which is flying with oneor more of a pitch, roll, or yaw, and which has a hard mounted radarsystem, introduces an error element in a determination of targetlocation, in body coordinates. As such radars typically operate withrespect to the line of flight, a coordinate system with respect to theline of flight has been developed and is sometimes referred to as adoppler coordinate system. FIG. 5 illustrates differences betweenaircraft coordinates and doppler coordinates. An x-axis of the dopplercoordinate system, Xd, is on the line of flight. A y-axis, Yd, and az-axis, Zd, at right angles to Xd, respectively are defined as acrossXd, and above and below Xd.

[0044] Therefore, if aircraft 2 is flying with no pitch, roll, or yaw,the body coordinate system aligns with the doppler coordinate system.For a positive roll, Xm and Xd are still aligned, while Yd rotates belowYm and Zd rotates to the left of Zm. For a positive yaw, Xd rotates tothe right of Xm, Yd rotates behind Ym, and Zd and Zm are aligned. For apositive pitch, Xd rotates above Xm, Yd aligns with Ym, and Zd rotatesahead of Zm. The complexity of having multiple of pitch, roll, and yaw,and determining a target position in aircraft body coordinates isapparent.

[0045]FIG. 6 is one embodiment of a doppler radar processing system 200.System 200 incorporates three radar antennas which receive reflectedradar pulses, the pulses having originated from a radar source. A leftantenna 202 receives the pulses and forwards the electrical signal toreceiver 204. Receiver 204 forwards the received radar signal to a dataacquisition unit 206. A right antenna 208 receives the pulses, at aslightly different time than left antenna 202, and forwards theelectrical signal to receiver 210. Receiver 210 forwards the receivedradar signal to a data acquisition unit 212. An ambiguity antenna 214also receives the reflected radar signal, and passes the received signalto a circulator 216. Circulator 216 functions to direct the transmitsignal to the antenna, and to direct the received signal from theantenna to receiver 220, thereby allowing a single antenna to be usedfor both transmitting and receiving. Receiver 220 forwards the receivedsignal to a data acquisition unit 222.

[0046] Data acquisition unit 206 provides a digital signalrepresentative of the signal received at left antenna 202 to a leftphase pre-processing unit 224. Similarly, representative signals arereceived at pre-processing units 226 and 228 from data acquisition units222 and 212, respectively. Data acquisition units 206, 212, and 222 areconfigured, in one embodiment, to sample received signals, and therebyreduce the data to a rate which allows a relatively low speed computerto process digitized radar data. In one embodiment, pre-processing units224, 226, and 228 perform a gate ranging function.

[0047] A phase processor 230 receives gated, filtered signals,representative of left, right, and ambiguity signals received at theantennas, and determines a phase relationship between each of the leftand ambiguous signal, the right and ambiguous signals, and the right andleft signals. The phase relationships between the signals are used,along with slant range, velocity and attitude readings in a phaseambiguity processing unit 232 to determine an interferometric angle to atarget. A body coordinate processor 233 utilizes the interferometricangle to determine an XYZ position of, for example, an aircraftemploying system 200 with respect to a current aircraft position,sometimes referred to herein as aircraft body coordinates.

[0048] A signal from data acquisition unit 222 is also received at anautomatic gain control (AGC) unit 234. A signal from AGC unit 234 ispassed to pre-processing units 236, 238, and 240. A filtered signal frompre-processing unit 236 is passed to range track processor 242 whichprovides a slant range signal to phase ambiguity processing unit 232 andaltitude information. Pre-processing unit 238 passes a filtered signalto a range verification processor 244. Pre-processing unit 240 passes afiltered signal to a range level processor 246, which also provides afeedback signal to AGC 234.

[0049]FIG. 7 is a block diagram of a digital processing section 300 forsystem 200 (shown in FIG. 6). Components in section 300, identical tocomponents of system 200, are identified in FIG. 7 using the samereference numerals as used in FIG. 6. Section 300 includespre-processing units 224, 226, 228, 236, 238, and 240 and processors230, 242, 244, and 246. Referring specifically to pre-processing units224, 226, 228, 236, 238, and 240, each includes a gate correlator 302, acorrelation band pass filter 304, a baseband I/Q mixer 306, and a swathband pass filter 308. A filter coefficients processor 309, in oneembodiment, is configured to provide at least a filter center frequencyin hertz, Fc, a filter bandwidth in hertz, B, and a filter samplingfrequency in hertz, Fs, to swath band pass filter 308, which uses Fc, B,and Fs in determination of filter coefficients. In one embodiment,processor 309 receives as input, an antenna mounting angle, velocityvectors in body coordinates, a pitch, and a slant range.

[0050]FIG. 8 is a block diagram of a correlation band pass filter 304(also shown in FIG. 7). An input signal 310, sometimes referred to asx(0), is fed into a summing element 312. An output of summing element312 is multiplied by a coefficient 313, which, in one embodiment has avalue of 1/K1 (further described below). After multiplication bycoefficient 313, an output signal 314, sometimes referred to as y(0), isgenerated. Another input into summing element 312 is provided by inputsignal 310 being delayed by a two sample delay element 316, whoseoutput, sometimes referred to as x(−2), is fed into summing element 312.Further, output signal 314 is fed back into a second two sample delayelement 318, whose output, sometimes referred to as y(−2), is multipliedby a second coefficient 319, and fed into summing element 312. In oneembodiment, coefficient 319 has a value of K3. Therefore, a presentoutput, y(0) is calculated as y(0)=(1/K1)x[x(0)−x(−2)]−(K2×y(−2)), whereK1=C+1, K3=C−1, K2=K3/K1, and C=1/Tan(π×bandwidth/f_(sample)) wherebandwidth and sample frequency are in hertz, and the angle for which thetangent is to be calculated is in radians.

[0051] In alternative embodiments, filter 304 is configured to filterrange ambiguity spectrum lines, filter out-of-band interference signalsand stretch the input signal, which is a pulse, to a continuous wave(CW) signal. Filter 304, in one embodiment, receives as input an outputof gate/correlator 302 (shown in FIG. 7) at a sample rate of 100 MHz, anIF frequency of 25 MHz, and has a bandwidth of 10 KHz. Therefore, inthis embodiment, there are four samples per IF frequency period.

[0052] A sample clock at 100 MHz provides samples at a 10 nsec rate. Forexample, a 4 μsec pulse repetition interval (PRI) (N=400 clocks per PRI)and two sample gate width, results in two non-zero gated return samples,x(0) and x(1), and 398 zero amplitude samples, x(2)−x(399), intocorrelation filter 304 during one PRI. In order to provide a filter ofreasonable processing size and speed, the zero amplitude samples whichdo not affect filter output are not processed by filter 304. Therefore,past outputs, for example y(−2), required in the filter feedbackconfiguration, as illustrated by delay elements 316 and 318, at the timeof non-zero inputs are not available. These past outputs are calculatedbased on filter outputs generated during and directly after the previousreturn (the previous non-zero samples), and filter droop characteristicsover a known pulse repetition interval.

[0053] In addition, one of the past outputs, y(−1), is not used becauseit has a feedback multiplier with a value of nearly zero in oneembodiment of filter 304, because of the narrow 10 kHz bandwidth.

[0054] In one exemplary embodiment, where F_(sample)=100 MHz, centerfrequency=25 MHz, and Bandwidth=8 KHz, coefficients are calculated asK1=3979.873661, K3=3977.873661, and K2=0.9994974715. Let P=the number ofsamples in a PRI. Filter 304 starts calculating at the beginning of agate width and continues for two counts after the end of the gate width.After the gate width +2 counts the next step is to calculate y(−2) andy(−1) and wait for x(P) data, the beginning of the next gate width,where x(P) is equivalent to x(0). Table 1 illustrates a generalprocedure for operation of filter 304, for low altitude radar data,track and phase gate of two sample widths, and a PRI of 400 μsec. Thecalculation for filter output y(0) requires filter output y(−2). Theexample of Table 2 example illustrates calculation of y(−2) where N=400,if PRI=4 μsec. TABLE 1 Correlation Filter Algorithm Example x(N) Count(N) Algorithm 0 397 y(−3) = y(397) 0 398 y(−2) = y(398) 0 399 y(−1) =y(399) x(0) 0 y(0) = (1/K1)[x(0) − x(−2)] − [K2 × y(−2)] x(1) 1 y(1) =(1/K1)[x(1) − x(−1)] − [K2 × y(−2)] 0 2 y(2) = (1/K1)[x(2) − x(0)] − ]K2× y(0)] 0 3 y(3) = (1/K1)[x(3) − x(1)] − [K2 × y(1)] 0 4 y(4) = 0 − K2 ×y(2) = −K2 × y(2) = (−K2)¹ × y(2) 0 5 y(5) = 0 − K2 × y(3) = −K2 × y(3)= (−K2)¹ × y(3) 0 6 y(6) = 0 − K2 × y(4) = −K2 × y(4) = −K2[(−K2) ×y(2)] = (−K2)² × y(2) 0 7 y(7) = 0 − K2 × y(5) = −K2 × y(5) = −K2[(−K2)× y(3)] = (−K2)² × y(3) 0 8 y(8) = 0 − K2 × y(6) = −K2 × y(6) =−K2[(−K2) × (−K2) × y(2)] = (−K2)³ × y(2) 0 9 y(9) = 0 − K2 × y(7) = −K2× y(7) = −K2[(−K2) × (−K2) × y(3)] = (−K2)³ × y(3) 0 10 y(10) = 0 − K2 ×y(8) = −K2 × y(8) = −K2[(−K2) × (−K2) × (−K2) × y(2)] = (−K2)⁴ × y(2) 011 y(11) = 0 − K2 × y(9) = −K2 × y(9) = −K2[(−K2) × (−K2) × (−K2) ×y(3)] = (−K2)⁴ × y(3)

[0055] In one embodiment, y(399) becomes y(0) if a range gate is movedin an inbound direction. The resulting P becomes 399. If a range gate ismoved in an outbound direction, y(1) becomes y(0), and the resulting Pbecomes 401. Algorithms shown for determination of y(4) through y(11)are used to formulate a general algorithm equation.

[0056] In addition to an example illustration of calculation of y(−2)with a P of 400 and a gate width of two clock counts, Table 2 alsoillustrates a general algorithm equation for counts (N) greater thanthree, (i.e. y(N)=(−K2)^(M)×y(2), for N even and y(N+1)=(−K2)^(M)×y(3),where M=(N(even)/2)−1. TABLE 2 General Algorithm Equation after N = 3Ein Count (N) Algorithm 0 396 y(−4) = (−K2)¹⁹⁷ × y(2) 0 397 y(−3) =(−K2)¹⁹⁷ × y(3) 0 398 y(−2) = (−K2)¹⁹⁸ × y(2) 0 399 y(−1) = (−K2)¹⁹⁸ ×y(3) x(0) 0 y(0) = (1/K1)[x(0) − x(−2)] − [K2 × y(−2)] x(1) 1 y(1) =(1/K1)[x(1) − x(−1)] − [K2 × y(−1)] 0 2 y(2) = (1/K1)[x(2) − x(0)] − [K2× y(0)] 0 3 y(3) = (1/K1)[x(3) − x(1)] − [K2 × y(1)]

[0057] In the embodiment described, for y(0) through y(3), the filteralgorithm is calculated because new x(N) and/or y(N) data are available.After the y(3) algorithm calculation, y(398) and y(399) are calculated,and the filter algorithm is configured to wait for x(400) data, wherex(400) is equivalent to x(0). If a range tracking algorithm dictatesthat x(0) be x(399), that is, the range gate causes the PRI to beshortened, then y(397) and y(398) are calculated. If the range trackingalgorithm dictates that x(0) be x(401), that is, the range gate causesthe PRI to be increased, then x(399) and x(400) are calculated. Thesignal phase is preserved by using the correct x(0) and y(−2). The PRIis not limited to 4 μsec and can have a wide range of values. The filteralgorithm is configured to set the N counter to count to 400 on the nextcycle unless the range tracking algorithm requires 399 or 401 counts. Ingeneral, a filter configured similarly to filter 304 is capable ofremoving up to about 95% of the mathematical operations that arerequired in known filter processing schemes.

[0058] Another exemplary embodiment of filter 304, for high altitudeoperation, incorporates a Barker code. Table 3 illustrates an exemplaryembodiment, with a chip width equal to four, a PRI of 4 μsec, and P=400.In the exemplary embodiment, a 13 bit Barker code is used, and inputsx(0) and x(1) are data, x(2) and x(3) are filled with zeros, x(4) andx(5) are data, x(6) and x(7) are filled with zeros, and the patterncontinues until N is equal to 51. Generally, the algorithm for N greaterthan 51 is given as y(N)=(−K2)^(M)×y(50), for N even, andy(N+1)=(−K₂)^(M)×y(51), where M=(N(even)−50)/2)−1. TABLE 3 Barker codesat high altitudes example x(N) Count (N) Algorithm 0 397 y(−3) = y(397)0 398 y(−2) = y(398) 0 399 y(−1) = y(399) x(0) 0 y(0) = (1/K1)[x(0) −x(−2)] −[K2 × y(−2)] x(1) 1 y(1) = (1/K1)[x(1) − x(−1)] −[K2 × y(−1)] 02 y(2) = (1/K1)[x(2) − x(0)] − [K2 × y(0)] 0 3 y(3) = (1/K1)[x(3) −x(1)] − [K2 × y(1)] x(4) 4 y(4) = (1/K1)[x(4) − x(2)] − [K2 × y(2)] x(5)5 y(5) = (1/K1)[x(5) − x(3)] − [K2 × y(3)] . . . . . . . . . 0 396 y(−4)= y(396) = (−K2)¹⁷² × y(50) 0 397 y(−3) = y(397) = (−K2)¹⁷² × y(51) 0398 y(−2) = y(398) = (−K2)¹⁷³ × y(50) 0 399 y(−1) = y(399) = (−K2)¹⁷³ ×y(51) x(0) 0 y(0) = (1/K1)[x(0) − x(−2)] − [K2 × y(−2)] x(1) 1 y(1) =(1/K1)[x(1) − x(−1)] − [K2 × y(−1)] 0 2 y(2) = (1/K1)[x(2) − x(0)] − [K2× y(0)]

[0059]FIG. 9 is a block diagram of a baseband IQ mixer 306. Mixer 306 isconfigured to reject negative Doppler shifts on the IF (IntermediateFrequency) input signal, which are behind aircraft 2, while allowing apositive doppler shift signal, from ahead of aircraft 2 to pass through.The positive doppler shift signal is equally forward as the negativedoppler shift signal is behind. Referring specifically to mixer 306, anIF in-phase portion includes a mixer 322 configured to operate at afrequency which is 1/PRI, where PRI is a radar pulse repetitioninterval, which converts the in-phase IF signal to Baseband (Doppler)frequency. Also included in the in-phase portion are a low pass filter324, a decimator 326, and an all pass filter 328. Referring specificallyto mixer 306, an IF quadrature portion includes a delay element 330,which produces the IF quadrature signal, and a mixer 332 configured tooperate at a frequency which is 1/PRI, where PRI is a radar pulserepetition interval, which converts the quadrature IF signal to Baseband(Doppler) frequency. Also included in the quadrature portion are a lowpass filter 334, a decimator 336, and an all pass filter 338. All passfilters 328 and 338 are configured to produce Baseband (Doppler)quadrature signals, which are received at a difference element 340,where the output of the all-pass filter 338 is subtracted from theoutput of the all-pass filter 328. The resulting difference signalcontains the positive or forward-looking Baseband (Doppler) signal,which is received at swath bandpass filter 308.

[0060] In particular embodiments, a frequency of data received at mixer306 is 25 MHz, and is referred to as an IF (intermediate frequency)signal. Mixer 306 in one embodiment, is configured to convert the 25 MHzIF signal to baseband (or Doppler) frequencies, and further configuredto reject negative Doppler frequencies. In specific embodiments, mixers322 and 332 are configured with PRIs which allow decimation of thesignal from correlation bandpass filter 304 to a 25 kHz sample rate.Specifically, in the embodiment shown, the allowed PRIs include 200,400, 500, 800, and 1000.

[0061] For purposes of description, a current input to low pass filter324 is given as x1(0). A current output of the low pass filter 324 isthen given as y1(0)=(1/K1)[x1(0)+x1(−1)]−[K2×y1(−1)], where x1(−1) andy1(−1) are respectively the previous input and output of the low passfilter 324. A current input to low pass filter 334 is given as x0(0). Acurrent output of the low pass filter 334 is then given asy0(0)=(1/K1)[x0(0)+x0(−1)]−[K2×y0(−1)], where x0(−1) and y0(−1) arerespectively the previous input and output of the low pass filter 334.K1 is 1+(1/tan(πfo/Fs2), and K2 is 1−(1/tan(πfo/Fs2), where fo isbandwidth and Fs2 is a sampling frequency of low pass filters 324 and334. In one embodiment, the sampling frequency of low pass filters 324and 334 is the received signal frequency, Fs1, of 100 MHz divided by thepulse repetition interval.

[0062] The signals output from low pass filters 324 and 334 are furtherdown sampled at decimators 326 and 336. In one embodiment, decimators326 and 336 are configured to sample at a frequency which is the pulserepetition interval multiplied by a sampling frequency, Fs3, of all passfilters 328 and 338, divided by the received signal frequency, or(PRI×Fs3)/Fs1.

[0063]FIG. 10 is a block diagram 350 of Baseband (Doppler) in-phaseall-pass filter 328 and Baseband (Doppler) quadrature all-pass filter338. In one embodiment, all-pass filter 328 and all-pass filter 338include four cascaded second-order infinite impulse response (IIR)filters, configured to generate Baseband (Doppler) quadrature signals.Referring specifically to all-pass filter 328, it includes filterelements 352, 354, 356, and 358, sometimes referred to herein as a, b,c, and d respectively. Referring to all-pass filter 338, it includesfilter elements 362, 364, 366, and 368, sometimes referred to herein ase, f, g, and h respectively.

[0064]FIG. 11 is a block diagram of one embodiment of a filter element380. Element 380 is a representation of all of filter elements 352, 354,356, 358, 362, 364, 366, and 368 (shown in FIG. 9). The followingdescription refers specifically to element 380, consisting of delayelements 392, 396, 400, 404, summing element 386, and gain elements 384,394, 398, 388, 402, 406. For the purposes of description the currentinput 382 is referred to as x(0). The current output 390 is then givenas y(0)=[(A0* x(0))+(A1*x(−1))+(A2*x(−2))−(B1*y(−1))−(B2*y(−2))]/B0,where x(−1) and y(−1) are respectively the previous input and output offilter element 380, and x(−2) and y(−2) are respectively theprevious-previous input and output of filter element 380. A0, A1, A2,B1, and B2 refer to the gain block coefficients.

[0065] In one specific embodiment, the above equation is applicable forall of filter elements 352, 354, 356, 358, 362, 364, 366, and 368 (shownin FIG. 9). The following are the coefficients for each filter element,the elements 352, 354, 356, 358, 362, 364, 366, and 368 beingrepresented by a, b, c, d, e, f, g, and h respectively, and BBfreq isthe base band sampling frequency, and T is I/BBfreq. In one embodiment,floating point precision is used.

[0066] Element a

[0067] a=1.0/0.3225;

[0068] w0=57.956;

[0069] A2=(4.0/T)/T+(2.0×w0×a/T)+w0×w0;

[0070] A1=(−8.0/T)/T+2.0×w0×w0;

[0071] A0=(4.0/T)/T−(2.0×w0×a/T)+w0×w0;

[0072] B2=(4.0/T)/T−(2.0×w0×a/T)+w0×w0;

[0073] B1=(−8.0/T)/T+2.0×w0×w0;

[0074] B0=(4.0/T)/T+(2.0×w0×a/T)+w0×w0;

[0075] Element b

[0076] b=1.0/0.4071;

[0077] w0=1198.2;

[0078] A2=(4.0/T)/T+(2.0×w0×b/T)+w0×w0;

[0079] A1=(−8.0/T)/T+2.0×w0×w0;

[0080] A0=(4.0/T)/T−(2.0×w0×b/T)+w0×w0;

[0081] B2=(4.0/T)/T−(2.0×w0×b/T)+w0×w0;

[0082] B1=(−8.0/T)/T+2.0×w0×w0;

[0083] B0=(4.0/T)/T+(2.0×w0×b/T)+w0×w0;

[0084] Element c

[0085] c=1.0/0.4073;

[0086] w0=16974.0;

[0087] A2=(4.0/T)/T+(2.0×w0×c/T)+w0×w0;

[0088] A1=(−8.0 T)/T+2.0×w0×w0;

[0089] A0=(4.0/T)/T−(2.0×w0×c/T)+w0×w0;

[0090] B2=(4.0/T)/T−(2.0×w0×c/T)+w0×w0;

[0091] B1=(−8.0/T)/T+2.0×w0×w0;

[0092] B0=(4.0/T)/T+(2.0×w0×c/T)+w0×w0;

[0093] Element d

[0094] d=1.0/0.3908;

[0095] w0=259583.5;

[0096] A2=(4.0/T)/T+(2.0×w0×d/T)+w0×w0;

[0097] A1=(−8.0/T)/T+2.0×w0×w0;

[0098] A0=(4.0/T)/T−(2.0×w0×d/T)+w0×w0;

[0099] B2=(4.0/T)/T−(2.0×w0×d/T)+w0×w0;

[0100] B1=(−8.0T)/T+2.0×w0×w0;

[0101] B0=(4.0/T)/T+(2.0×w0×d/T)+w0×w0;

[0102] Element e

[0103] e=1.0/0.3908;

[0104] w0=152.05;

[0105] A2=(4.0/T)/T+(2.0×w0×e/T)+w0×w0;

[0106] A1=(−8.0/T)/T+2.0×w0×w0;

[0107] A0=(4.0/T)/T−(2.0×w0×e/T)+w0×w0;

[0108] B2=(4.0/T)/T−(2.0×w0×e/T)+w0×w0;

[0109] B1=(−8.0/T)/T+2.0×w0×w0;

[0110] B0=(4.0/T)/T+(2.0×w0×e/T)+w0×w0;

[0111] Element f

[0112] f=1.0/0.4073;

[0113] w0=2326.03;

[0114] A2=(4.0/T)/T+(2.0×w0×f/T)°w0×w0;

[0115] A1=(−8.0/T)/T+2.0×w0×w0;

[0116] A0=(4.0/T)/T−(2.0×w0×f/T)+w0×w0;

[0117] B2=(4.0/T)/T−(2.0×w0×FT)°w0×w0;

[0118] B1=(−8.0/T)/T+2.0×w0×w0;

[0119] B0=(4.0/T)/T+(2.0×w0×f/T)°w0×w0;

[0120] Element g

[0121] g=1.0/0.4071;

[0122] w0=32949.65;

[0123] A2=(4.0/T)/T+(2.0×w0×g/T)°w0×w0;

[0124] A1=(−8.0/T)/T+2.0×w0×w0;

[0125] A0=(4.0/T)/T−(2.0×w0×g/T)+w0×w0;

[0126] B2=(4.0/T)/T−(2.0×w0×g/T)°w0×w0;

[0127] B1=(−8.0/T)/T+2.0×w0×w0;

[0128] B0=(4.0/T)/T+(2.0×w0×g/T)°w0×w0;

[0129] Element h

[0130] h=1.0/0.3225;

[0131] w0=681178.9;

[0132] A2=(4.0/T)/T+(2.0×w0×h/T)+w0×w0;

[0133] A1=(−8.0/T)/T+2.0×w0×w0;

[0134] A0=(4.0/T)/T−(2.0×w0×h/T)°w0×w0;

[0135] B2=(4.0/T)/T−(2.0×w0×h/T)°w0×w0;

[0136] B1=(−8.0/T)/T+2.0×w0×w0;

[0137] B0=(4.0/T)/T+(2.0×w0×h/T)°w0×w0;

[0138]FIG. 12 is a block diagram of one embodiment of a swath band passfilter 308. Filter 308 is a first order band pass filter which iscentered on the doppler frequency. Filter 308 receives as input asignal, En, output from IQ mixer 306 (shown in FIG. 9). Further inputsinclude a filter center frequency in hertz, Fc, a filter bandwidth inhertz, B, and a filter sampling frequency in hertz, Fs, which areprovided.

[0139] A filtered output signal, Eo, is determined according toEo=(A0/B0)×En−(A0/B0)×En×Z⁻²−(B1/B0)×Eo×Z⁻¹−(B2/B0)×Eo×Z⁻². Referringspecifically to filter 308, the input signal, En 422 is received andmultiplied by a coefficient 424, with a value of A0/B0, and then appliedto a summing element 426. The output of summing element 426 is filteroutput 428. Input 422 is also delayed two counts by a two sample delayelement 430 whose output is multiplied by coefficient 432, with a valueof −A0/B0, and then applied to summing element 432.

[0140] Output 428 is multiplied by a sample delay element 434, whoseoutput is multiplied by a coefficient 436, with a value of −B1/B0, andthen applied to summing element 432. Output 428 is also multiplied by atwo sample delay element 438, whose output is multiplied by acoefficient 444, with a value of −B2/B0, and then applied to summingelement 432. Coefficients for filter 308 are determined according toWb=2πB, which is bandwidth in radians, Wu=2π×(Fc+B/2), which is an upper3 db point of filter 308 in radians, and W1=2π×(Fc−B/2), which is alower 3 db point of filter 308 in radians. The coefficient A0 is2×Fs×Wb, B0 is (4×Fs²)+(2×Fs×Wb)+(W1×Wu), B1 is (2×W1×Wu)−(8×Fs²), andB2 (4×Fs²)−(2×Fs×W1)+(W1×Wu).

[0141]FIG. 13 is a block diagram of a filter coefficients processor 309(also shown in FIG. 7) which, in one embodiment, is configured toprovide inputs to swath band pass filters 308 (shown in FIGS. 7 and 12).Processor 309 is configured to provide center frequencies Fc, for rangeswaths and phase swaths, and filter bandwidths, B, in hertz, for trackand phase swaths and level and verify swaths. By controlling swathfilter center frequencies, processor 309 is able to keep the dopplerswath centered in the antenna beam. Also filter bandwidth is controlled.The filter bandwidth is directly related to a down track swath width onthe ground such that a charge time for filter 308, inversely butdirectly related to bandwidth, is equal to the time it takes aircraft 2to fly across the swath width. Therefore, filter bandwidth is matched tovelocity of aircraft 2, and requires minimal processing. By knowing theantenna mounting angle, and the pitch of the aircraft, an angle to theantenna beam center is known, as described below, and a center frequencyis calculated, generally, according to Fc=2×Velocity×sin (angle)/radarwavelength.

[0142] Referring specifically to processor 309, an antenna mountingangle and velocity vectors in body coordinates are input to determine adoppler velocity, Vr 460, at a range swath center frequency according toVr=Vv×Cos(90−r−a)=Vv×Sin(a+r), where Vv=(Vx²+Vz²)^(0.5), whereVx=velocity component on body x axis and Vz=velocity component on body zaxis, a=ATan(Vz/Vx), and r is the antenna mounting angle. A range swathcenter frequency, Fr 462 is determined according to Fr=2×Vr/L, where Lis a wavelength, and in one specific embodiment, is 0.2291 feet. Avelocity component on body y axis, Vy, is not used to center swath inantenna beam as the component has a value of zero since the antenna isfixed to a y axis of the body.

[0143] Processor 309 is also configured to determine a phase swathdoppler velocity, Vp 464, which is delayed behind the range swath by atime equal to the range processing delay. Vp is calculated asVp=Vv×Cos(90−(r−p)−a)=Vv×Sin(a+r−p), where Vv=(Vx²+Vz²)^(0.5), whereVx=velocity component on body x axis and Vz=velocity component on body zaxis, a=ATan(Vz/Vx), r is the antenna mounting angle, andp=(T×Vx/H)×(180/π) in degrees, where T=1/πB and is a delay through rangeswath filter, T×Vx is vehicle movement on body X axis, B is the swathbandwidth, and H is altitude in feet. Phase swath center frequency 466is calculated according to Fp=2×Vp/L, where L is a wavelength, and inone specific embodiment, is 0.2291 feet.

[0144] Processor 309 is configured to determine a track and phase swathbandwidth, B 468 according to B=Vx/(0.6(H)^(0.5)) in hertz, where H isaltitude in feet. A level and verify swath bandwidth 470 is calculatedas a ratio of level and verify bandwidths to track and phase bandwidths,K, multiplied by track and phase swath bandwidth 468. FIG. 14 is avector diagram 500 which illustrates the calculations above described.In one embodiment, if the radar is in a range search mode, search rangeinstead of altitude is used to calculate bandwidth.

[0145] Together, filters 308 and processor 309 automatically configurethe radar doppler filter center frequency and bandwidth to achievebetter radar performance over varying terrain and varying aircraftaltitude, roll, and pitch than known systems. The determined centerfrequency operates to maintain the radar swath at an approximate centerof the antenna beam. The calculated bandwidth is a bandwidth thatcontrols the track swath width on the ground, and is calculated suchthat the filter time constant is equal to the time it takes the vehicleto move a corresponding swath width distance. The bandwidth correspondsto a time over the target and provides information as to how long asecond swath lags a first swath. Phase channel swaths are set behind inposition to account for a processing time of range processor 242 (shownin FIG. 7). The calculations of center frequency and bandwidth provide amechanism for keeping a swath slightly in front of the aircraft suchthat a positive doppler shift is realized.

[0146]FIG. 15 is a block diagram of a phase processor 230 (also shown inFIGS. 6 and 7). Phase processor 230 includes three phase detectors 510,512, and 514. In one embodiment, phase detectors 510, 512, and 514 areconfigured with an input and a reference input, and further configuredto determine a phase difference between the input and the referenceinput. Phase processor 230 is configured to receive processed radarreturn data, from swath band pass filters 308 (shown in FIG. 7), asdescribed above, for all of a left channel, a right channel, and anambiguous channel. Determination of phase difference in return data forthe three channels allows for an accurate position determination for anobject from which radar data was returned.

[0147] In the embodiment shown, phase detector 510 is configured toreceive ambiguous channel return data as input, with left channel returndata as a reference, and further configured to determine and output aphase difference between the left and ambiguous channels. Phase detector512 is configured to receive right channel return data as input, withambiguous channel return data as a reference, and further configured todetermine and output a phase difference between the ambiguous and rightchannels. Phase detector 514 is configured to receive right channelreturn data as input, with left channel return data as a reference, andfurther configured to determine and output a phase difference betweenthe left and right channels.

[0148]FIG. 16 is a block diagram of phase detector 510 (shown in FIG.15). Phase detectors 512 and 514 are of the same configuration. Phasedetector 510 incorporates a plurality of in-phase all pass filters 328and quadrature all pass filters 338 (shown above in FIGS. 9 and 10).Specifically, an input is received at a first in-phase filter 520(AP1.1) and a first quadrature filter 522 (AP1.2). A reference input isreceived at a second in-phase filter 524 (AP2.1) and a second quadraturefilter 526 (AP2.2). A multiplier 532 is configured to multiply outputsfrom filters 520 and 526. Another multiplier 534 is configured tomultiply outputs from filters 522 and 524. A third multiplier 536 isconfigured to multiply outputs from filters 520 and 524. A fourthmultiplier 538 is configured to multiply outputs from filters 522 and526. An output of multiplier 534 is subtracted from an output ofmultiplier 532 with a subtraction element 540 which produces a Y output542. An output of multiplier 536 is added to an output of multiplier 538with an addition element 544 which produces an X output 546. Aprocessing element 548 is configured to determine an arctangent of Youtput 542 divided by X output 546, which is the phase difference, inradians, between the input and the reference input.

[0149] In mathematical form, Y output 542 is calculated asY=(AP1.1×AP2.2)−(AP1.2×AP2.1), X output 546 is calculated asX=(AP1.1×AP2.1)+(AP1.2×AP2.2), and the phase difference is ATAN (Y/X).

[0150] In one embodiment, in-phase filters 520 and 524 and quadraturefilters 522 and 526 include the four cascaded second order infiniteimpulse response (IIR) filters as described in FIG. 10. Further, in theembodiment, filters 520 and 524 are configured to include in-phasefilter elements 352, 354, 356, and 358, (shown in FIG. 10) and areconfigured with coefficients which correspond to elements a, b, c, and drespectively as described above. Referring to quadrature filters 522 and526, they are configured to include quadrature filter elements 362, 364,366, and 368, (shown in FIG. 10) and are configured with coefficientswhich correspond to elements e, f, g, and h respectively as describedabove.

[0151] Once phase differences between the right, left, and ambiguouschannels has been determined, as described above, the phase differencesare used, in one embodiment, to determine and interferometric angle tothe target. FIG. 17 is a block diagram of phase ambiguity processingunit 232 (also shown in FIG. 6). In one embodiment, phase ambiguityprocessing unit 232 is configured to receive an electrical phasedifference between the ambiguous channel and the left radar channel fromphase detector 510, an electrical phase difference between the rightchannel and the ambiguous radar channel from phase detector 512, and anelectrical phase difference between the right channel and the left radarchannel from phase detector 514.

[0152] Phase ambiguity processing unit 232 includes a phase bias adjustunit 570 which provides a phase shift value which compensates for phaseshifts which occur in the routing of the radar signals, from receipt atan antenna and through cabling and processing areas within aircraft 2.It is accepted that most phase shifting of signals occurs due to cablingfor the routing of signals. Phase bias adjust 570 compensates for theambiguous channel with respect to the left radar channel. Phase biasadjust 572 compensates for the right channel with respect to theambiguous radar channel. Phase bias adjust 574 compensates for the rightchannel with respect to the left radar channel.

[0153] The compensated phase difference signals are received at a phaseambiguity resolver 576. In one embodiment, phase ambiguity resolver 576is implemented using software, and determines a physical(interferometric) angle to a target which originally reflected the radarsignals received. Phase ambiguity resolution is further described below.After resolution of phase ambiguous signals, the physical angle signalis filtered utilizing a low-pass filter 578, and an angular position ofthe target with respect to aircraft body coordinates (X,Y,Z) isdetermined from the physical angle to the target using body coordinatesprocessor 233 (further described below). The determined position, in oneembodiment, is 90 degrees minus a half angle of a cone whose axis is aY-axis of the body of aircraft 2. The target is on the cone surface,therefore providing the subtraction from 90 degrees above described.TABLE 4 Phase Ambiguity Resolution Matrix θ_(LA) θ¹ = θ_(LA) θ¹ =(θ_(LA) − 360) θ¹ = (θ_(LA) + 360) Φ = sin⁻¹(θ¹/K1) Φ = sin⁻¹(θ¹/K1) Φ =sin⁻¹(θ¹/K1) θ_(AR) θ¹ = θ_(AR) θ¹ = (θ_(AR) − 720) θ¹ = (θ_(AR) − 360)θ¹ = (θ_(AR) + 360) θ¹ = (θ_(AR) + 360) Φ = sin⁻¹(θ¹/K2) Φ =sin⁻¹(θ¹/K2) Φ = sin⁻¹(θ¹/K2) Φ = sin⁻¹(θ¹/K2) Φ = sin⁻¹(θ¹/K2) θ_(LR)θ¹ = θ_(LR) θ¹ = (θ_(LR) − 720) θ¹ = (θ_(LR) − 360) θ¹ = (θ_(LR) + 360)θ¹ = (θ_(LR) + 360) Φ = sin⁻¹(θ¹/K3) Φ = sin⁻¹(θ¹/K3) Φ = sin⁻¹(θ¹/K3) Φ= sin⁻¹(θ¹/K3) Φ = sin⁻¹(θ¹/K3) θ_(LR) θ¹ = (θ_(LR) − 1080) θ¹ =(θ_(LR) + 1080) Φ = sin⁻¹(θ¹/K3) Φ = sin⁻¹(θ¹/K3)

[0154] Table 4 is a phase ambiguity resolution matrix which is utilized,in one embodiment, to determine a physical angle to a target based uponelectrical phase differences. A calculated electrical angle phasedifference, θ, is equivalent to [(360×S)/λ]×sin(Φ) or K×sin(Φ), where Φis the physical angle of the target in aircraft coordinates, S is aseparation between the two antenna elements in feet, and λ is awavelength of the radar signal in feet. In one particular embodiment,separation between the left antenna and the ambiguous antenna is 0.2917feet (3.5 inches), separation between the ambiguous antenna and theright antenna is 0.7083 feet (8.5 inches), and the separation betweenthe left antenna and the right antenna is 1 foot (12 inches). In theembodiment, the wavelength of the radar is 0.2291 feet. Therefore, inthe embodiment, and referring to Table 4, K1 is (360×0.2917)/0.2291, orabout 458.4, K2 is (360×0.7083)/0.2291, or about 1113.25, and K2 is(360×1)/0.2291, or about 1571.64. Physical angles are then determinedaccording to Φ=sin⁻¹(θ/K).

[0155] As antenna separation, radar wavelength, and aircraft positionmay all affect a timing of radar signals received at the variousantennas, phase differences, which are determined as described above,will change at varying rates. In the embodiment illustrated in Table 4,physical angles are calculated for multiple electrical phasedifferences, and the true physical angle is a solution which providesapproximately the same physical angle calculation, in each of the threerows (within a couple of degrees). Using the first antenna pairing (leftand ambiguous), and based on antenna separation, three possible physicalangles are determined from the electrical phase difference received fromphase detector 510. As the second antenna pairing (ambiguous and right)are further apart, five possible physical angles are determined. Thelast antenna pairing (left and right) are the furthest apart, thereforeseven possible physical angles are determined. As described above, oneof the physical angles from each group of physical angle calculations,will be roughly equivalent, thereby providing an unambiguous physicalangle solution. In such a system it is important to note that separationin antenna pairing cannot be a multiple of radar wavelength.

[0156]FIG. 18 is a chart 600 illustrating varying electrical phasedifferences between three antenna pairings. Chart 600 helps toillustrate the process above described. As varying electrical phasedifferences between the three antenna pairings are charted, a singlemechanical (physical) angle can be determined from the varyingelectrical phase difference plots for each antenna pairing. That is, fora physical angle, there is one solution which provides a phasedifference for each radar channel grouping which is approximatelyequivalent to the calculated phase differences for the channelgroupings.

[0157]FIG. 19 is a block diagram which illustrates inputs to and outputsfrom body coordinate processor 233 (also shown in FIG. 6). Processorreceives the phase detector angle to the target from phase ambiguityresolver 576 via low pass filter 578 (described above in FIG. 17).Processor 233 further receives the doppler swath filter centerfrequency, and the filter bandwidth, a range to the target in feet, andvelocity in pitch, roll and azimuth. Utilizing the processing describedbelow, processor 233 is configured to determine a distance to the targetin aircraft body coordinates. In one embodiment, the distance isdetermined in feet for aircraft body coordinates x, y, and z. Processor233 further determines a velocity with respect to aircraft bodycoordinates in x and z.

[0158]FIG. 20 is a detailed block diagram of body coordinate processor233 of FIG. 19. Target range, vehicle velocity in pitch, roll, andazimuth, plus the swath filter center frequency and bandwidth are inputinto a doppler circle equation processor 620, which is configured todetermine doppler circle equations. The circle is determined using theswath filter center frequency equation Fc=[2×V×cos(β)]/L, where V isvelocity, L is wavelength, and β is an angle with respect to a line offlight, which is determined through manipulation of the above equation.Therefore, β=cos⁻¹((Fc×L)/(2×V)). A radius of the doppler circle, Rd, iscalculated according to Rd=target range×sin (β). A distance of thedoppler circle, Xd, from the aircraft is determined according toXd=target range×cos(β). FIG. 21 is provided to illustrate the equationswith regard to the doppler circle as derived above.

[0159] An example calculation is used to further illustrate. Inputs todoppler circle equation processor 620 include a range to target of 2000feet, a velocity of 800 feet/second, a wavelength of 0.229 feet, and adoppler swath filter center frequency of 1213 Hertz. The angle withrespect to the aircraft line of flight, β, is determined asb=cos⁻¹((1213×0.229)/(2×800))=80 degrees. The doppler circle radius, Rd,is 2000×sin(80)=1969 feet, and distance of the doppler circle, Xd, is2000×cos(80)=347 feet.

[0160] Again referring to FIG. 20, processor 233 further includes aninterferometric circle equation processor 622 which is configured todetermine interferometric circle equations in body coordinates.Processor 622 receives as input a target range and the interferometricangle (or phase detector angle), a, to the target as calculated by phaseambiguity resolver 576 (shown in FIG. 17). An interferometric circleradius, Ri, is calculated as Ri=target range×cos(a). A location of theinterferometric circle on a Ym axis is determined as Ym=targetrange×sin(a). Referring to the example above, and including aninterferometric angle input of 15 degrees, the radius of theinterferometric circle, Ri, is 2000×cos(15), or 1932 feet. The locationof the circle on the Ym axis, Ym is 2000×sin(15), or 518 feet. FIG. 22is provided to illustrate the equations with regard to theinterferometric circle as derived above.

[0161] Again referring to FIG. 20, a doppler to body coordinatetransformation processor 624 within processor 233 uses the dopplercircle equation, and pitch, roll, and yaw inputs to transform thedoppler circle into body coordinates. Finally, at intersection processor626 which is configured to solve equations to determine an intersectionof the interferometric circle equation with the doppler circle equationthat has been transformed into body coordinates.

[0162] In one embodiment, transforming begins by a determination of avelocity vector in body coordinates, from navigation data, N, (in pitch,roll, and yaw) according to ${{{\begin{matrix}V_{X}^{N} \\V_{Y}^{N} \\V_{Z}^{N}\end{matrix}}{{{TRANSPOSE}\quad {MATRIX}}}} = {\begin{matrix}V_{X}^{BODY} \\V_{Y}^{BODY} \\V_{Z}^{BODY}\end{matrix}}}\quad,$

[0163] where the transpose matrix is given by ${\begin{matrix}{{\cos (\psi)}{\cos (\theta)}} & {{- {\sin (\psi)}}{\cos (\theta)}} & {\sin (\theta)} \\{{{\cos (\psi)}{\sin (\theta)}{\sin (\varphi)}} - {{\sin (\psi)}{\cos (\varphi)}}} & {{{- {\sin (\psi)}}{\sin (\theta)}{\sin (\varphi)}} - {{\cos (\psi)}{\cos (\varphi)}}} & {{- {\cos (\theta)}}{\sin (\varphi)}} \\{{{\cos (\psi)}{\sin (\theta)}{\sin (\varphi)}} + {{\sin (\psi)}{\sin (\varphi)}}} & {{{\cos (\psi)}{\sin (\varphi)}} - {{\sin (\psi)}{\sin (\theta)}{\sin (\varphi)}}} & {{- {\cos (\theta)}}{\cos (\varphi)}}\end{matrix}}\quad,{{and}\quad \psi \quad {is}\quad {azimuth}},{\theta \quad {is}\quad {pitch}\quad {and}\quad \varphi \quad {is}\quad {{roll}\quad.}}$

[0164] Velocity unit vectors (direction cosines) are given in bodycoordinates as a_(x)=V_(x)(V_(x) ²+V_(y) ²+V_(x) ²)^(1/2),a_(y)=V_(y)(V_(x) ²+V_(y) ²+V_(z) ²)^(1/2), and a_(z)=V_(z)/(V_(x)²+V_(y) ²+V_(z) ²)^(1/2).

[0165] Intersection processor 626 is configured to determine bodycoordinates which are calculated as X₁=D×a_(x), Y₁=D×a_(y), Z₁=D×a_(z),where the velocity vector D, is given as R×cos(β), andβ=cos⁻¹(Fc×L/2×V). B is the doppler cone angle, Fc is the swath filtercenter frequency, R is the range to the target, V is (V_(x) ²+V_(y)²+V_(z) ²)^(1/2), and L is the wavelength of the radar.

[0166] A position of the target in body coordinates is also calculatedby intersection processor 626 as y=R×sin(A), where A=measured phaseangle in body coordinates. The coordinate z is calculated asz=(−b±(b²−4ac)^(1/2))/(2×a), where a=1+(Z₁/K₁)², b=(−4Z₁×KT/(2X₁)²), andc=(KT/2X₁)²−KA. KA is calculated as (R×cos(A))², KB is calculated as(R×sin(B))², KY=(y−Y₁)², and KT is calculated as KT=KA+KY−KB+X₁ ²+Z₁ ².The coordinate x is calculated according to x=(KA−z²)^(1/2).

[0167] While determining a position of a radar target with respect to,for example, an aircraft body, as described in detail above isnecessary, it is also necessary in certain application to determine arange to a target. As is well known, in high altitude radar operations,it is possible that multiple radar transmit pulses will be transmittedbefore a return pulse is received. This is sometimes referred to as theambiguous radar range problem. FIG. 23 illustrates one solution to theproblem, the solution being to modulate radar transmit pulses 650 with aphase code. Implementation of the code, which involves a phase shiftingof individual pulses of radar transmit pulses 650, allows asynchronization of transmit pulses 650 with return pulses 652 which arereceived by a radar. Synchronization of the phase encoded radar pulseswith the returned pulses is sometimes referred to as correlation.

[0168] In one embodiment, correlation is accomplished by implementationof a encoded radar scheme, and by looking for deviations in the returnpulses from a reference, or starting altitude. FIG. 24 is a blockdiagram illustrating inputs to and outputs from range verificationprocessor 244 (also shown in FIGS. 6 and 7). In one embodiment,verification processor 244 is configured to step through encoded returnsignals and determine a main lobe of the return signal to determine arange to, for example, a target.

[0169] Verification processor 244 is configured to receive as inputs, adetected radar return, which has been gated and demodulated.Verification processor 244 also receives as input a present internalrange to the target, and a command from the radar search logic to be ineither of a search mode or an acquisition mode. Verification processor244 is configured with a variable mainlobe threshold factor (describedbelow) and a verification dwell time, which is the time processor 244 isallocated to determine if an amplitude of a return signal exceeds thethreshold factor. A verify status output is set true of the amplitude ofthe radar return exceeds the threshold value, thereby signifying thatthe transmit radar pulses and return radar pulses are correlated. If notcorrelated, the verify status output is false, and processor 244provides a corrected range position to range processor 242 (shown inFIG. 7).

[0170]FIG. 25 is a flowchart 670 illustrating one embodiment of anautocorrelation process performed by processor 244. Referring toflowchart 670, a verify gate is set 672 to an internal range, from oneof track or search. It is then determined whether a radar return isacquired 674 from within a verify gate, the gate attempting to align thechips of transmitted and received codes. If no target is acquired 674,then processor 244 is configured to return to reset the verify gate. Ifa target is acquired 674, then an amplitude of the return is determined676. In addition, the threshold factor is set to, for example, fourtimes the determined amplitude and a counter is set to zero. The verifygate is stepped 678 out one chip of the code, the counter isincremented, and a dwell time passes before an amplitude of a return isagain read. If the amplitude read is determined 680 not to be above thethreshold factor, the counter is checked 682. If the counter isdetermined to be less than one less than the number of chips within thebarker code, the verify gate is again stepped 678, and the steps arerepeated, until the threshold factor is exceeded or the counter is equalto one less than the number of chips within the code. In one exemplaryembodiment, a thirteen bit code is used, therefore the counter has amaximum value of twelve. In one embodiment barker codes are used forencoding the radar signals.

[0171] If the threshold factor is not exceeded, the original acquisitionis an acquisition on the main lobe of the return, and the transmit andreturn codes are aligned, and the internal range as determined byprocessor 244 is correct, resulting in a verification status being set684 to verify.

[0172] If the threshold factor is exceeded, then the transmit and returncodes have become aligned. If the internal range has been moved 686 morethan two range gates, the process illustrated by flowchart 670 beginsanew. If there is a less than two range gate movement 686, the searchlogic of the radar is set 688 to not verify, and is moved by the valueof the counter, in order to align the transmit and receive barker codes.The process illustrated by flowchart 670 again begins. The continuousprocessing of encoded radar transmit and return signals by processor,provides a favorable solution to the known radar range ambiguity problemby constantly stepping through the codes to ensure receipt of anunambiguous radar range return.

[0173] In one embodiment, the above described verification processingfor radar range ambiguity is applied continuously during flight, notjust during initial acquisition. In utilization of such a system, theverification processing is applied in order to resolve range ambiguityduring acquisition, but the processing is continuously applied afteracquisition, throughout the flight. The continuous processing is done inorder to ensure that if the transmit and received pulses becomemisaligned (loose correlation) the misalignment will both detected andcorrected. Loss of correlation could occur due to, for example, a rangediscontinuity due to severe aircraft rolls or a sudden change in terrain(i.e. flying over a cliff).

[0174] The verification processing is further illustrated through anexample.

[0175] In one embodiment, a phase code is used to resolve radar rangeambiguities and particularly a 13 bit phase code provides 20×log(13) or22 dB of rejection to range sidelobes. However, if verificationprocessor 244 should, for some reason, line itself on an ambiguous sidelobe, even if the mainlobe is for example 22 dB higher in amplitude,verification processor 244 will stay aligned with the sidelobe as longas there is a greater than 22 dB sensitivity margin. As stated above,one such example is flying over a sharp and deep cliff where a maximumradar track rate is less than a rate at which the range changes over thecliff. However, in practice, and assuming an ambiguous range sidelobe islined up, a transition to a decreased sensitivity margin will normallyresult in a less than sufficient margin to track the ambiguous rangeside lobe. Examples include flying over poor reflectivity ground orencountering a severe aircraft roll. The result is verificationprocessor 244 realigning into a proper and unambiguous line up onto themain lobe. Thus an ambiguous radar range does, after some time, normallycorrect itself. However, and especially with auto pilot systems, severeand dangerous aircraft altitude corrections will result during the timeof this very undesirable ambiguous range condition.

[0176] The method illustrated in flowchart 670 resolves the aboveillustrated situation by continuously searching for the main lobe, whiletracking what is believed to be the correct position, or lobe. If duringthe ambiguity processing, or verification background search, it isdetermined that an ambiguous range is being tracked, an immediatecorrection is made to get the radar onto the correct range (i.e. themain lobe). To detect if the radar is on an ambiguous range track, the20LogN equation is utilized to continuously determine differencesbetween the main lobe, and undesired side lobes.

[0177] The above described methods and systems describe a digital signalprocessing solution to known radar target position and range ambiguityproblems. Use of digital signal processing techniques therefore enablesa radar system to perform faster and more accurate airborne processingthan known radar ambiguity solutions. While the invention has beendescribed in terms of various specific embodiments, those skilled in theart will recognize that the invention can be practiced with modificationwithin the spirit and scope of the claims.

What is claimed is:
 1. A method for processing radar return data todetermine a physical angle, in aircraft body coordinates to a target,the radar return data including a phase difference between radar returndata received at an ambiguous radar channel and a left radar channel, aphase difference between radar return data received at a right radarchannel and an ambiguous radar channel, and a phase difference betweenradar return data received at a right radar channel and a left radarchannel, said method comprising: adjusting a phase bias for the threephase differences; resolving phase ambiguities between the three phasedifferences to provide a signal; and filtering the signal to provide aphysical angle to the target in aircraft body coordinates.
 2. A methodaccording to claim 1 wherein resolving phase ambiguities comprises:determining a plurality of physical angle solutions for each receivedphase difference; and determining an unambiguous physical angle basedupon physical angle solutions which are approximately equal from eachreceived phase difference.
 3. A method according to claim 2 whereindetermining a plurality of physical angle solutions for each receivedphase difference comprises: determining physical angle solutionsaccording to Φ=sin⁻¹(θ¹/K1), where θ¹, is determined as θ¹=θ_(LA),θ¹=(θ_(LA)−360), and θ¹=(θ_(LA)+360), K1 is [(360×S_(LA))/λ], whereS_(LA) is a separation between a left antenna element and an ambiguousantenna element in feet, λ is a wavelength of the radar signal in feet,and θ_(LA) is a received electrical phase angle difference between aleft radar channel and an ambiguous radar channel; determining physicalangle solutions according to Φ=sin⁻¹(θ¹/K2), where θ¹, is determined asθ¹=θ_(AR), θ¹=(θ_(AR)−720), θ¹=(θ_(AR)−360), θ¹=(θ_(AR)+360), andθ¹=(θ_(AR)+720), K2 is [(360×S_(AR))/λ], where S_(AR) is a separationbetween an ambiguous antenna element and a right antenna element infeet, λ is a wavelength of the radar signal in feet, and θ_(AR) is areceived electrical phase angle difference between an ambiguous radarchannel and a right radar channel; and determining physical anglesolutions according to Φ=sin⁻¹(θ¹/K3), where θ¹, is determined asθ¹=θ_(LR), θ¹=(θ_(LR)−1080), θ¹=(θ_(LR)−720), θ¹=(θ_(LR)−360),θ¹=(θ_(LR)+360), θ¹=(θ_(LR)+720), and θ¹=(θ_(LR)+1080), K3 is[(360×S_(LR))/λ], where S_(LR) is a separation between a left antennaelement and a right antenna element in feet, λ is a wavelength of theradar signal in feet, and θ_(LR) is a received electrical phase angledifference between a left radar channel and a right radar channel.
 4. Amethod according to claim 3 wherein S_(LA) is about 0.2917 feet, S_(AR)is about 0.7083 feet, S_(LR) is about one foot, and λ is about 0.2291feet.
 5. A processor configured to: resolve phase ambiguities betweenmultiple received phase difference signals; and determine a physicalangle in aircraft body coordinates to a target based upon the resolvedphase ambiguities, the phase difference signals having been determinedbased upon radar return data received at each of an ambiguous radarchannel, a left radar channel, and a right radar channel.
 6. A processoraccording to claim 5 wherein to resolve phase ambiguities, saidprocessor is configured to determine a plurality of physical anglesolutions for each received phase difference.
 7. A processor accordingto claim 6 wherein to determine a physical angle in aircraft bodycoordinates to a target said processor is configured to determine whichphysical angle solutions provide an unambiguous physical angle to thetarget.
 8. A processor according to claim 7 wherein the unambiguousphysical angle is an angle which is a solution for at least one of thephase angle solutions for each received phase difference.
 9. A processoraccording to claim 6 wherein said processor is configured to determinethe plurality of phase angle solutions according to: Φ=sin⁻¹(θ¹/K1),where θ¹, is determined as θ¹=θ_(LA), θ¹=(θ_(LA)−360), andθ¹=(θ_(LA)+360), K1 is [(360×S_(LA))/λ], where S_(LA) is a separationbetween a left antenna element and an ambiguous antenna element in feet,λ is a wavelength of the radar signal in feet, and θ_(LA) is a receivedelectrical phase angle difference between a left radar channel and anambiguous radar channel; Φ=sin⁻¹(θ¹/K2), where θ¹, is determined asθ¹=θ_(AR), θ¹=(θ_(AR)−720), θ¹=(θ_(AR)−360), θ¹=(θ_(AR)+360), andθ¹=(θ_(AR)+720), K2 is [(360×S_(AR))/λ], where S_(AR) is a separationbetween an ambiguous antenna element and a right antenna element infeet, λ is a wavelength of the radar signal in feet, and θ_(AR) is areceived electrical phase angle difference between an ambiguous radarchannel and a right radar channel; and Φ=sin⁻¹(θ¹/K3), where θ¹, isdetermined as θ¹=θ_(LR), θ¹=(θ_(LR)−1080), θ¹=(θ_(LR)−720),θ¹=(θ_(LR)−360), θ¹=(θ_(LR)+360), θ¹=(θ_(LR)+720), and θ¹=(θ_(LR)+1080),K3 is [(360×S_(LR))/λ], where S_(LR) is a separation between a leftantenna element and a right antenna element in feet, λ is a wavelengthof the radar signal in feet, and θ_(LR) is a received electrical phaseangle difference between a left radar channel and a right radar channel.10. A radar signal processing circuit comprising: a radar gatecorrelation circuit configured to sample radar return data from left,right, and ambiguous radar channels at a sampling rate; a correlationbass pass filter configured to stretch the sampled radar return data toa continuous wave (CW) signal; a mixer configured to down sample anin-phase component and a quadrature component of the CW signal to adoppler frequency; a band pass filter centered on the doppler frequency;a phase processor configured to receive processed radar return data fromsaid band pass filter, said phase processor further configured todetermine a phase difference between radar return data from an ambiguouschannel and a left channel, a phase difference between radar return datafrom an right channel and the ambiguous channel, and a phase differencebetween radar return data from the right channel and the left channel;and a processing unit configured to receive the three phase differences,adjust a phase bias for the three phase differences, resolve phaseambiguities between the three phase differences to provide a signal, andfiltering the signal to provide a physical angle to a target in aircraftbody coordinates.
 11. A radar signal processing circuit according toclaim 10 wherein said processing unit is configured to resolve phaseambiguities by determining a plurality of physical angle solutions foreach received phase difference.
 12. A radar signal processing circuitaccording to claim 11 wherein to provide a physical angle in aircraftbody coordinates to a target said processing unit is configured todetermine which physical angle solutions provide an unambiguous physicalangle to the target.
 13. A radar signal processing circuit according toclaim 12 wherein said processing unit configured to determine anunambiguous physical angle which is an angle that provides a solutionfor at least one of the phase angle solutions for each received phasedifference.
 14. A radar signal processing circuit according to claim 11wherein said processing unit is configured to determine a plurality ofphysical angle solutions for each received phase difference according toΦ=sin⁻¹(θ¹/K1), where θ¹, is determined as θ¹=θ_(LA), θ¹=(θ_(LA)−360),and θ¹=(θ_(LA)+360), K1 is [(360×S_(LA))/λ], where S_(LA) is aseparation between a left antenna element and an ambiguous antennaelement in feet, λ is a wavelength of the radar signal in feet, andθ_(LA) is a received electrical phase angle difference between a leftradar channel and an ambiguous radar channel; Φ=sin⁻¹(θ¹/K2), where θ¹,is determined as θ¹=θ_(AR), θ¹=(θ_(AR)−720), θ¹=(θ_(AR)−360),θ¹=(θ_(AR)+360), and θ¹=(θ_(AR)+720), K2 is [(360×S_(AR))/X], whereS_(AR) is a separation between an ambiguous antenna element and a rightantenna element in feet, λ is a wavelength of the radar signal in feet,and θ_(AR) is a received electrical phase angle difference between anambiguous radar channel and a right radar channel; and Φ=sin⁻¹(θ¹/K3),where θ¹, is determined as θ¹=θ_(LR), θ¹=(θ_(LR)−1080), θ¹=(θ_(LR)−720),θ¹=(θ_(LR)−360), θ¹=(θ_(LR)+360), θ¹=(θ_(LR)+720), and θ¹=(θ_(LR)+1080),K3 is [(360×S_(LR))/λ], where S_(LR) is a separation between a leftantenna element and a right antenna element in feet, λ is a wavelengthof the radar signal in feet, and θ_(LR) is a received electrical phaseangle difference between a left radar channel and a right radar channel.15. A radar signal processing circuit according to claim 14 wherein saidprocessing unit is configured an S_(LA) of about 0.2917 feet, an S_(AR)of about 0.7083 feet, an S_(LR) of about one foot, and λ of about 0.2291feet.